Electronics – Page 6 – gerrysweeney.com

Fully Programmable Modular Bench Power Supply – Part 9

By gerrysweeney on September 9, 2012July 10, 20179 Comments

After a fairly disappointing start with the 12-bit DAC from Microchip I have chnaged the DAC to a 16-bit resolution part LTC2602 from Linear Technology. The goal of getting a resolution that is as good as the HP3631A Power Supply means getting a programmable resolution of 1mv for the low voltage range and 10mv for the high voltage range. The circuit below achieves these goals with a high degree of accuracy and no missing codes throughout the range. The design also exceeds the dynamic voltage and current range of the HP shown in the table below. Furthermore I made a statement that I would aim to not only achieve the same resolution but I would also attempt to improve on it by a factor of 10, that means getting a resolution of 100µV on the low voltage range and 1mV on the high voltage range. The 16 bit resolution of the DAC was not quite enough to achieve this, there were missing codes so I used the DAC modulation technique I used for the 12-bit DAC and extended the 16-bit DAC to 18 bits of resolution which allowed me to achieve the ten-fold programming resolution with no missing codes across the entire range – marvellous 🙂

Here is a brief video overview of the modified hardware and test environment

And here is a video demonstration of programming, measuring and the achieved resolution.

I focused some more on noise measurements and found that modulating the output of the 16-bit DAC created no detectable ripple – at least with the test equipment that I have. In fact the noise I am measuring swamps any ripple that is being created now. Interestingly, the digital signals from the micro controller driving the DAC (the SPI bus) is now creating more noise than the ripple effect of switching between two adjacent codes. Because the digital switching does impose some noise, there is a compromise to be had – you can get higher resolution but suffer some additional noise. For low-noise applications you can forego the additional resolution and get better signal to noise performance. Thankfully this can be controlled entirely in software so can be easily made a programmable option which is what I will do.

Of course this is just the control circuitry, I have no idea how the noise or accuracy will be represented through the linear regulator part of the design in practice – we shall see.

I have broken the schematic into several pages to make it easer to read.

I am happy that the control circuitry and the firmware is now good enough for a Version 1 baseline – this is what we have now. I have increased each range to show the possible extent of each gain option. In the final design the power and voltage ranges required and programmed in the UI/front panel will dictate the actual upper limits of each range, but this table shows the extent of what is possible.

Volt Range	Volt Resolution	Volt Steps	Current Range	Current Resolution	Current Steps	Regulator Gain
0-8v	1mv/1µV	8000/80000	0-8A	1mA/0.1mA	8000/80000	64W max	x2
0-20V	10mV/1mV	2000/20000	0-20A	10mA/1mA	2000/20000	400W max	x5
0-40V	10mV/1mV	4000/40000	0-4A	10mA/1mA	4000/40000	160W max	x10

In Part 10 I am going to create the first PCB for the project – Whoo-hoo. However, instead of designing the entire PSU printed circuit board at this stage I am going to focus on the digital control circuitry with the ADC, DAC and Analog switch only. I am going to create a small PCB as a module with a simple 0.1″ pin header (Arduino shield style) with the ADC, DAC, Analog switch, PIC micro controller and power supply parts so I can do the first cut of a board layout to make sure the grounding and noise issues are addressed. It’s not my intention to make this part of the final PSU module but it will be a useful module for prototyping other systems and of course I will not waste too much time if I get it totally wrong :/

Catch you next time….

Fully Programmable Modular Bench Power Supply – Part 8

By gerrysweeney on August 17, 2012September 4, 202219 Comments

After my initial tests there are two things I now know for sure. The first is I cannot treat the DAC like a programmable voltage source, it’s not going to be accurate enough and secondly, because of the INL of the DAC the resolution needs to be high enough to give me headroom to “trim” the DAC to the correct voltage. On this basis there really are only two practical options to achieve the goal.

The first option is to use a mapping table. Assuming I have a DAC with enough resolution it would be feasible to build some form of calibration rig which could step through each desired voltage then trim the code written to the DAC to get as close as possible for the resolution of the DAC and store that number in a table which can then be written to micro controller as a lookup table. The problem with this approach is its messy, and it would mean that if the calibration was lost in use it would need the calibration rig to re-calibrate it. This is not an ideal solution.

The second option, and the one I will take forward is to build a system where I have an ADC and a DAC and the system is configured so I can read the DAC output. When a given voltage it programmed the ideal DAC code is written to get within a few millivolts of the desired output; and then a software loop is used to trim the DAC code up or down as required to get as close to the required output voltage as possible. If the PSU is to measure up (no pun intended) then it needs a metering function regardless so all I need to do is design the system so I can switch the ADC’s inputs to the DAC for the calibration cycle.

The ADC needs a good resolution, it needs to be accurate and it needs to be low cost. The conversion speed is also important, if its too slow the calibration cycle would take too long to complete. I found a good part for this in the form of the LTC2402 Delta Sigma ADC device from Linear Technologies. I must say at this point that Linear Technologies design support is really great, they have been kind enough to furnish me with sample devices which I have been testing in the design. I have also tested a couple Analog Devices parts too and they were also kind enough to furnish me with sample devices, and these were very close in performance and a bit cheaper too (so not excluded just yet) but for performance in this application Linear Technologies parts so far have the edge – I will talk more about that later on and in future articles because I have tested a few different DAC’s and ADC’s. The following schematic shows the digital control circuitry which is extended to include the ADC chip and a 74HC4053D analogue switch which is used to switch the input of the ADC from monitoring the PSU regulator outputs to monitoring the outputs of the DAC for calibration read-back.

One of the interesting attributes of the ADC chip is it’s a two channel ADC, but in reality though it is actually a single channel ADC with a two channel multiplex on the front end so it samples each channel alternately. This made for an interesting problem to solve because the calibration loop needs the conversions from the ADC to be as fast as possible. The chip its self can only perform a maximum of 6.2 samples per second according to the data sheet, but because of the input multiplexer it can only actually achieve 3.1 samples per second per channel. For normal monitoring three updates per second is acceptable but for calibration I wanted better performance. I configured the analogue switch so in calibration mode I can use both inputs of the ADC to monitor a single channel of the DAC which gives me a calibration loop speed of 6.2 trim and read back per second.

Unlike the DAC, the INL of the ADC is significantly better mainly due to the nature of how it works. The architecture is a switch capacitor system which unlike laser cut resistor strings the charging and discharging of current into a capacitor is significantly more predictable and linear.

https://www.youtube.com/watch?v=f-0m5m7rUQQ
PLEASE NOTE: The LED flash is continuous but the video frame rate make it looks like it stopes and starts – which is does not.

In software I have tired two different calibration schemes. The first is the simplest, it sets the ideal code for the target voltage, measures the output of the DAC and does a calculation to determine the difference between the ideal code and the actual code required, then applies this. The second scheme is more complicated. It is the same as the first scheme but then with a second phase of micro trimming which essentially sets a window around the centre point and then adjusts the codes in steps re-reading the output; you can think of this as an electronic/software version of someone turning a variable resistor while watching a meter to get a precise voltage. The second scheme also takes more time as it needs to perform multiple read-backs in order to complete the trim cycle. I have limited the trim cycle to a maximum of eight iterations, in practice it seems to take between 2 and 7 most of the time.

(P.S. The command ‘pr’ means “property read”, it reads the EEPROM configuration of the module)

I re-tested with the spot voltages in the previous blog post and the bottom line is, 14-bits is not enough. The MCP4922 is great for the $$ but its not good enough for this project so I will not bother creating another spot voltage table for this variation, that’s a waste of time given we have established the DAC is just not up to the job. In the next article I will try out two 16-bit DAC chips, one from Analog Devices and the other from Linear Technology.

Thanks and keep watching….

Microcontroller Source Code

Schematic and PCB Software – I Wish!

By gerrysweeney on July 17, 2012July 30, 20125 Comments

A long time frustration of mine has been the lack of good quality, easy to use electronic design software, I have tried a few and I am even a licensed user of the tool I consider to be the best of a bunch which is Diptrace (http://www.diptrace.com/), but its poor enough in some areas that even after buying a licence I am left with a burning desire to look for better options or as I have been sometimes known to do – write my own!

There are many packages available but all have their problems, usability, quality and intuitiveness are all lacking. Most of the packages are long-standing code bases that have been around for years and all appear dated in use. Packages are not interoperable and are based on proprietary file and library formats so once you are using one you are essentially tied in. You are also mainly limited to Windows with very limited support for Mac and Linux. The top of the range packages are outrageously expensive and closed off to the hobbyist/small business/one-man designer operations, this market is served by a number of budget systems, the best of which is still really only average. The more expensive solutions are packed full of features that are rarely used by most, Altium Designer (http://www.altium.com/) is one such tool and its users complain a lot about the lack of R&D dollars spent on the core functionality as they see it, instead Altium put their effort into other things like integrated FPGA design capability or integrated software development features – all a nice idea but not really desirable when its to the detriment of features on the core product capabilities that most people use on a daily basis (Disclaimer: this view of Altium is not my own, it’s what I have read/picked up from various forums when I was seriously considering buying a license for the tool).

I would love to see an affordable EDA tool that would serve both hobby/maker community as well as professional users. The ideal package would avoid needless bloat and complexity by specifically focusing only on the four main functional areas that actually help electronics engineers design, document and make electronic projects. I would like this tool to be fast, intuitive to use (I mean intuitive in terms of windows/mac UI users as opposed to making it work like 90’s AutoCAD) and works equally as well on Windows, Mac and Linux. The four key areas in order of importance in order of priority are: –

Component Library Manager & Content

At the heart of any good EDA tool lies a component library manager and a component library (actual content, not just a nice editor). Almost every tool out there suffers from poor quality library content or tools to make creating library content easy. Most tools require the users to build their own libraries and components despite there being a finite number of standard footprints and symbols. There is of course always a need to create other specific components which is not to be ignored but the most common component symbols, footprints and components should be provided in the library by default. Perhaps a nice-to-have but some kind of central on-line component library database that would allow community contribution and sharing of components really easily right from within the software its self should exist. I believe that in the absence of a decent library of components a not even a great tool is any good.

Component Editor

The ability to create components easily is critical, If it’s a common component it should be a simple matter of finding it on a global database and importing it into your local library. However, when the component is not available it must be a trivial matter to create a new one. Most components are simple block components represented on a schematic diagram as a rectangle with pins extending from any of its four sides. There are no vector graphics drawn in these components so they can easily be defined in data form rather than in graphic/plot form as many products from the 80/90’s seem to do. Creating such components should be really fast to do and require very little layout design, simple data entry and a bit of layout to finish up is all that should be needed. The other kind of component is a vector graphic component, for example a transistor or diode. The best way of thinking of these components is to imagine a block component described above with the relevant pins added and a vector graphic drawn inside the rectangle — and the rectangle outline being hidden. It would seem sensible to allow the import of standard scalable vector graphics (such as SVG files) rather than try to re-create a graphics design tool within the system which many other solutions do – but very badly. There are good, free tools that are infinitely better than anything that could be made and maintained so to reflect the desire not to bloat the system with features that are rarely used then just importing such graphics would be a better starting point.

Schematic Capture

All electronics projects/designs start with a schematic diagram. In my ideal package, the schematic editor would be the heart of the entire solution that would drive many of the design aspects – this is different to most other tools which would appear to put PCB board design at the heart of the solution – this I think is a historical thing. A schematic diagram is a drawing that shows how all of the electronic components in a project connect to each other electrically (not physically). A good schematic capture tool should be easy to use first and foremost and should support all of the key features like hierarchical and multi-sheet designs, in-schematic component pin re-organisation to suit schematic being designed, smooth interactive zooming, panning and publish-quality rendering and printing. Net list creation should be automatic and make sense to the user. A decent Bill of Materials (BOM) really ought to be generated from the design with no effort, and not just a stupid text file or excel spread sheet with limited information. Native elements of a schematic should be implemented as first class items, and not simply as another type of component which most packages seem to do. Most of all designing a schematic should be a pleasant activity, the presentation of the schematic should be beautiful, smooth graphics, good use of subtle colours and tones and different stroke wights to add character to the components and their visualisation; I say this because I am opposed to the blocky lines, and ugly component presentation that most packages seem to provide. Just like any other creative activity, electronic design is a form or art and must be nice to look at. There is absolutely no reason with today’s computing capability available to use why you cannot create better graphics in designer mode.

PCB Design

Many electronics projects lead to a PCB design. A PCB design tool should be easy to use and feature packed, most tools do this reasonably well because before fully integrated solutions were available PCB design tools were the primary focus. As a result most tools today have their roots in PCB design with schematic capture playing second fiddle. It’s not unheard of for people to use two different vendors packages, one for PCB design and the other for schematic capture. There are standard file interchange formats for net-lists which makes this possible. Easy and intuitive to use, smooth zooming, panning and track editing, easy access to layer controls and component placement/organisation are all key. Manual routing should be supported and made as easy as possible, despite autoroutermania there is still very much a need for manual layout/editing.

PCB Auto Routing

Slightly controversial I imagine is my thoughts on auto-routing, it would appear on the face of it that one could get significantly better and faster results with more powerful computers in a cluster. Auto routing could be done by uploading the PCB landmass and net list to a cloud based service that would carry out the actual routing retiring the route design back to the tool. Not sure how well this approach would be received, but I certainly like the idea of it.

2D Panel Design

Most other tools do not seem include this feature which is odd because after PCB design it’s very common to want to make basic panels with holes, cut-outs and shapes etc. There are many generic 2D packages out there but my feeling is there ought to be something that is specifically designed for electronics related designs such as front/back panels, chassis panels and flat parts etc. Like the component library a cut-out and shapes library here would be a fantastic productivity boost in creating panels.

Is this too much too ask? Is it a pipe dream? If you have a view/opinion I would love to hear it, please take the time to comment.

Fully Programmable Modular Bench Power Supply – Part 6

By gerrysweeney on July 14, 2012July 30, 20121 Comment

In Part 5 I talked about the target ranges that the PSU module should be able to support and expanded in detail on the voltage range calculations, and I said the next part would be looking at the DAC control circuitry. However, I really need to do the same exercise for current limit control in order to understand the required current control voltage range and resolution we need to ensure our DAC selection is appropriate for our current limit control too.

Current sensing starts with a “current shunt” which is simply a low value resistor in series with the load inside the power supply that will drop a small amount of voltage across it that is directly proportional to the current flowing through it. Selecting the right value is critical to ensure that your current sensing circuit operates in the correct range while making sure that the power dropped across the shunt resistor does not take the component outside of a reasonable thermal range, ideally you want your shunt resistor to run at ambient room temperature under the largest supported load. The PSU design calls for current handling of up to 5A. Just like in the DAC calculations we want to aim for nice round numbers so we pick a resistor value that is going to meet all of these requirements. In the design I have chosen a value of 0.1 ohms which if you do a simple OHMS LAW calculation “E = I / R” you will see that for a 0.1R resistor we get 500mV dropped across it when 5A of current is flowing. The amount of power dissipated in the shunt resistor at 5A can be determined by the formula “P = E x I” which works out to be 2.5W so we need to make sure that our chosen resistor is able to deal with that power and adequately dissipate the two and a half watts of heat generated.

A 0.1R resistor is a common high-power resistor value, and for 5/10W resistors they are typically a wire wound construction. However, the tolerance on these resistors is around the 5-10% mark which we need to take into consideration. Just like the potential divider in the voltage sense circuitry, the current sense resistor is the only component in the circuitry that has a direct bearing on the accuracy of the the current sensing, so a 5% tolerance is really not good enough for what we want. Unfortunately, as soon as you need precision at low resistor values things start to get very expensive. Think about this, if you want a 0.1R resistor at 1% precision then our tolerance needs to be +/- 0.01 ohms which is very hard to make, hence the cost. I do wan’t the precision but without the cost, and am happy to be within 1% for the current sense, so a nice solution is to use 10 x 1R 1% 250mW resistors in parallel to create a 2.5w 0.1R 1% resistor. These resistors are cheap at just a couple of cents each and the tolerance will be tight enough that any error can be easily trimmed out in software calibration. The prototype I built uses through-hole parts that are soldered in but mounted about 5mm above the surface of the board providing airflow around them to help keep them cool. In production I may opt for using SMT resistors and a larger copper area to help dissipate the heat. Here is a photo of the construction described.

If accuracy beyond 1% is important you can hand pick resistors from different manufacturers batches and try to match the resultant restore created. If you fall slightly above the desired resistance you can trim it down by adding higher values in parallel. Or you can go down the “buy an expensive precision shunt resistor route”. In testing my own shunt though, I found I was within a few miliohms of 0.1R which is about what I would expect for 1% tolerance resistors. When measuring these resistances you need a meter that has a Relative or Null mode because you need to compensate for the meter leads and the mechanical contact. I used decent fluke leads and the supplied heavy duty croc clips, I used two different meters and nulled on the croc clips being shorted together before measuring the resistor. Here are the results…

Measured with a Fluke 289 on the low ohms range

Measured with a HP34401A on the ohms range

Now we know that our current sense resistor is good this will give us 0 to 500mV for 0-5A of current we need to sense that voltage and scale it up so we can do something useful with it. The MAX4080F I talked about in Part 4 does both of these things for us. This device is a purpose made high-side current sense amplifier and precision x5 gain amplifier with a buffered output. Being a high-side device its powered from the +V rail so we don’t need to do anything special. In our configuraiton and the x5 gain of the MAX4080F part we now have a buffered output voltage in the range of 0-2.5v for the current range of 0-5A. This control voltage is fed to the constant current error amp on the negative input creating a control loop that prevents the regulator from delivering any more current than is programmed. The current is set by applying the same voltage on the + input of the current error amp as the voltage that will be seen on the output of the MAX4080F device. If we provide a reference voltage of 500mV then the regulator will limit output current to 1A – Nice…

I thought it worth covering the two main schemes you can use to measure DC current in a power supply when using a current shunt resistor – these are known as high-side and low-side sensing.

High side sensing is done on the hot regulated rail, often before the regulator pass device, whereas low-side sensing is done on the ground return path from the load. I have used high-side current sensing in this design because its the significantly better of the two schemes for a number of reasons, and its simple to implement because the MAX4080F makes it so. In the absence of such a useful single low cost purpose made device, then it could be tempting to go with low-side sensing because of simplicity. However, low-side current sensing designs introduce problems that our design will probably not tolerate. In low-side sensing the current sense resistor is placed in the load’s return path to ground, this has the undesirable effect of disturbing the common ground of the overall system when under load which can lead to stability problems, especially if we are doing remote sensing. Even worse than that is the risk that in a lab setup you are using a ground referenced PSU and in your test setup you ground your negative terminal of the PSU that has low-side current sensing which will effectively bypass or at the very least degrate the current sensing capability of your PSU which could cause a lot of damage, especially if your short circuit protection depends on the current sensing. Another problem with low-side sensing is as the current increases the ground reference and sense point will move apart as more voltage is dropped across the current sense resistor, this will introduce offset errors in voltage sensing. Low-side sensing is easy to do and easy to measure with a cheap op-amp but high-side current sensing brings so many advantages that $2 for the MAX4080F or another device like is money well spent I think.

One minor problem with the MAX4080F for this design is the common mode range of the device, the data sheet says it only goes down to 4.5V which is likely to be a problem in the case of a short circuit on the PSU output, really the common mode range needs to be able to extend to zero. I will need to test further to determine if it will be a problem in this design and possibly identify an alternative device – if anyone knows of any that are worth looking at please add a comment.

So back to the PSU, a quick recap on the ranges we are woking towards but this time with a focus on current sense and control.

Range	Current Range	Current Set Resolution	Full Scale Sense voltage	Volts per Milliamp	Current Steps
A	0-5A	1mA	2.5V	500µV	5000
B & C	0-2A	1mA	1V	500µV	2000
D	0-1A	1mA	500mV	500µV	1000

As you can see from the table above we have a more limited control range requirement so one configuration covers all ranges. However, there is a slight problem inasmuch as the design only gives us 2.5v out of the high-side amp at full scale, which is only 500µV per step which is half of what we get from our DAC at 1mV per step with a 4.096v reference. I can divide down the DAC output by x2 but then I fall short of the target range of 5A so I am going to have to rely on increasing the DAC’s resolution by modulating the DAC output – only measuring this physically will verify if thats good enough. Now we have a good sense of the control ranges need, in Part 7 I will work up a circuit, get a micro controller talking to the DAC chip and do some some perormance measurements to see what we come up with…. I know I said that in Part 5, but this time I mean it 🙂

Fully Programmable Modular Bench Power Supply – Part 5

By gerrysweeney on July 11, 2012July 14, 20121 Comment

I have set the regulator aside for a while, its time to turn to the control circuitry. However, before we get into the circuitry we need to do some pretty important calculations to help us select the right parts for the job. If you recall one of the aims for the power supply module is to be able to set its personality (supported voltage and current range) to suit a variety of voltage ranges. I have chosen a number of ranges to work within and specified them in the table below. Obviously its a relatively trivial task to add other ranges in the future so long as we stay within the confines of the working ranges of the components we have used in the system.

Range	Volt Range	Volt Resolution	Volt Steps	Current Range	Current Resolution	Current Steps	Approximate Power	Regulator Gain
A	0-6v	1mV	6000	0-5A	1mA	5000	30W	x2
B	0-16V	5mV	3200	0-2A	1mA	2000	40W	x5
C	0-20V	5mV	4000	0-2A	1mA	2000	40W	x10
D	0-30V	10mV	3000	0-1A	1mA	1000	30W	x10

The values in the table above are what we use to establish the correct control circuitry so they probably deserve a bit of explanation. The voltage and current range columns speak for themselves, they tell us what the expected programmable range of the regulator will be. The voltage and current resolution columns tell us the size of each step in the range, so for example in range “A” which has a resolution of 1mv/1mA we can program the output volts from 0V to 6V in 1mV steps, and for the current we can program the limit from 0A to 5A in 1mA steps. The volts and current steps columns tell us how many digital codes are needed to represent the desired output voltage and resolution.

Getting our Voltage under Control

In oder to get a digitally controlled voltage we need to use a DAC (Digital to Analog Converter). A DAC takes a digital “code” and translates this directly to a static voltage. However, DAC’s typically operate in the 3.3v to 5.5v range so to control higher voltage ranges we need to ensure that the regulator circuit has gain. This is achieved by simply dividing the measured output voltage using two precision resistors before feeding back into the error amplifier. The column called “Regulator Gain” shows the gain factor required for the specified voltage range. You will see in the Part 4 schematic the gain has been set to x10 by R4 and R5. It is also with noting that these two resistors are the only two components in the entire regulator circuit that have any impact on the accuracy of the programmed output. The effects of inaccuracies in the other components are automatically and fully compensated for by the DC Servo effect created by the control loop. It is critical that R4 and R5 are accurate (I am using 0.1% tolerance resistors) and also have good temperature stability.

Alright, get to the point, what about the DAC!

In order to select the right DAC resolution and voltage reference (more on that later) we need to establish the range and resolution of the required control voltage. This is important because we must include the gain of the regulator which in turns affects the resolution requirements of the DAC. The following table expands on this with a key values we need to consider.

Range	Output Range	Resolution	Steps	Regulator Gain	Reference Range	Reference Resolution
A	0-6v	1mV	6000	x2	0-3v	500µV
B	0-16V	5mV	3200	x5	0-3.2v	1mV
C	0-20V	5mV	4000	x10	0-2V	500µV
D	0-30V	10mV	3000	x10	0-3V	1mV

We calculate the control voltage range by dividing the target voltage by the gain multiplier value. For example, on range A, in order to obtain an output voltage of 6V we need a control voltage of 3V, calculated as 6 ÷ 3 = 2. To work out the reference control voltage resolution required we simply take the maximum controlling reference voltage needed and divide it by the number of steps required, in this case its 500µV per step.

From the above table we can clearly see that we need a maximum of 6000 steps to cover all ranges, which requires a DAC resolution of at least 13-bits. That is really inconvenient because DAC’s increase in price as the bits go up and there is a very notable jump in price as you go above 12-bits. A potential solution to this problem is to increase the step size on the 0-6v range from 1mV to 2mV which would half the resolution but would bring me nicely into range for a 12-bit DAC but I did not want to do that, its not what I set out to do, I needed to maintain the target resolution of 1mV on the 6v range. Thinking about it some more I thought that I could probably increase the resolution of the DAC in software using a technique whereby the DAC output is continuously set to a series of values at high speed to create an average voltage that can sit between between steps, this is known as dithering or modulating the DAC output. The downside to doing this is you introduce a peak ripple which reflects the continuously changing voltage level. However, filtering this out a straightforward affair because the amplitude of the ripple created is actually very small, so as long as we have enough current drive from the DAC a simple second-order passive low-pass filter is all that should be needed. Something worth noting though – the amplitude of the ripple voltage as an overall percentage of the output voltage is not linear; at lower voltages the ripple constitutes a larger proportion of the overall signal and therefore introduces more error. In practice though this should still end up well within tolerance, and if not we can always put in a higher resolution DAC!

One final thing is we need to decide on a reference voltage to use. DAC’s require a reference voltage in order to provide an accurate output voltage. Think of a DAC as a programmable potentiometer where the output is the wiper, the reference voltage is strapped across the two outer pins and the code sets the position of the potentiometer. If we have a 12-bit DAC that gives us 4,096 steps so if we use a reference voltage of exactly 4.096V we will have a theoretical output of 0-4.096V in 1mV steps which are nice round numbers that make software easier to develop, so that’s what we will go for.

So in summary, we have a design plan, we have our calculations and we know we need at least a 12-bit DAC. I am going to start with a DAC from Microchip MCP4822 which is a dual 12-bit DAC, its low cost and seems a reasonable starting point to do some experimenting, it also has a built in 2.048v reference and a x2 amplifier giving us the 0-4.096v @ 1mV steps.

In Part 6 I will work up a circuit, get a micro controller talking to the DAC chip and do some some perormance measurements to see what we come up with.

Low-cost Temperature Measurement

By gerrysweeney on July 1, 2012August 16, 20167 Comments

As part of the Fully Programmable Modular Bench Power Supply project I needed to measure the temperature of the heat sink in order to provide some thermal protection. The aim is simple enough, monitor the temperature of the heat sink and if it gets too hot shut it down until the temperature returns to an acceptable level. As it turns out there is more to this than one might think.

Firstly I needed to decide on a sensing device. There are numerous options to choose from falling into three main categories of passive, active linear and active digital.

Passive devices like RTD’s and Thermister’s are available in all sorts of types and are basically resistors that change value with temperature. These devices need circuitry around them and can be complicated to use mainly because their rate of change is expressed as a percentage per degree which in practice means a very non-linear response curve to temperature change. For my needs this was too complicated, at best I would need some kind of signal conditioning, feed into the micro controlled ADC and a software map of temperature vs. voltage to get a reading.

Digital devices such at the Dallas DS1821 can be connected to a micro-controller or in the case of the DS1821 a 1-wire serial bus. The device can be queried and it will return information about the current temperature it is sensing. These are very nice all digital devices but carry the penalty of cost, they are not cheap, costing upwards of $2 a piece. Because of cost I decided not to pursue the use of these

Analog devices are solid state devices that provide a variable voltage out that directly relates to temperature being sensed and are basically linear. These devices are typically low cost, simple three pin devices with power in, ground and volts out. The following are good devices to look at: –

I was thinking about the Bill of Materials for my project and was not really wanting to add yet another part. I had read somewhere that silicone has a very linear temperature co-efficient of around -2mV/°C of temperature change, which means if you measure the voltage drop across a diode or transistor junction you will see a 2mv decrease in voltage dropped for every degree of temperature rise. As my circuit was already using some PN2222 devices which are general purpose NPN transistors and LM358‘s I thought it would be an interesting exercise to see if a simple discrete temperature sensor could be built.

Pen, paper, breadboard, some bits and an hour yielded the following circuit:

This circuit provides a voltage swing of +1.5v to +3.5v for a temperature range of -50°C to +150°C. I was only able to test it down to a few degrees below zero but it tracked all the way up to 150°C without any problems. The data sheet for the PN2222 states it will operate down to -50°C so I have no doubt it will track downwards. Room temperature was at around 2v, I used a type K thermocouple and a Fluke 289 meter to calibrate it. The results of this circuit are surprisingly good and I did not add one single item to the BOM on the project, every component was already in use elsewhere in the circuit.

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Unfortunately the problem came when I needed to couple the device thermally. The main heat generating device on the project in question is an IRF540 in a TO-220 package. Bonding the PN2222 to the heat sink close to the device was my plan but in testing what I found was the lag between the IRF540 junction temperature and the heat skink temperature was enough that under extreme conditions the device will self-destruct long before the heat skink is hot enough to tell the control circuitry to shutdown. I found that the temperature on the plastic front of the device was much hotter than on the heat sink so I think the sensor really ought to somehow be mounted on the front of the device where the thermals are closer to the junction temperature than the heat sink. This exercise also convinced me to change the device to a bigger IRFP240 which is physically bigger and has better thermal properties. Although the data sheet for the IRF540 puts the device well within spec, I am more comfortable with the bigger package, more surface area must mean better thermal properties.

Back to the BOM – the PN2222’s that are in use in the project are in an SMT package so there was no practical way to thermally couple the two devices which meant I would need to add a TO-92 packaged device to the BOM and that really took away any advantage I had gained by using components already in the BOM for the project. I thought it still worth publishing the circuit as it may be of some use to others, pretty much any op amp and any silicon diode or transistor will yield much the same results.

In the end I opted to use the MCP9700A device from Microchip Technology, this is accurate enough for what I need and costs around $0.50 for a one-off. Still more expensive than the discrete design but I am trading that off for simplicity. The MCP9700A has a known temperature range and a known voltage output so interfacing to the MCU is as simple as connecting it to one of the ADC channels and measuring 10mV per degree of temperature – easy…..